Communication semiconductor integrated circuit device for use in a mobile communication device for correcting variations in the oscillation frequency of a transmission oscillator by calibrating a current of the charge pump

ABSTRACT

The invention provides a communication semiconductor integrated circuit (RF IC) that, when a transmission oscillator is incorporated into a semiconductor chip, secures the oscillation operation over a wide frequency range, prevents a deterioration of a transmission spectrum, and thereby enhances the accuracy of an oscillation frequency. The integrated circuit corrects a dispersion of the KV characteristic of the transmission oscillator by calibrating a current Icp of the charge pump inside the phase control loop. More in concrete, the integrated circuit measures a KV value Kv of the transmission oscillator, and calibrates the current Icp of the charge pump so that Kv·Icp falls into a predetermined value.

CROSS-REFERENCE TO RELATED APPLICATION

The present application is a continuation of application Ser. No.10/862,626, filed Jun. 8, 2004, now U.S. Pat. No. 7,187,911; whichclaims priority from Japanese Application Patent No. JP 2003-323995,filed Sep. 17, 2003, the content of which is incorporated by reference.

BACKGROUND OF THE INVENTION

The present invention relates to a technique effective in an applicationto incorporating a voltage controlled oscillator (VCO) capable ofswitching an oscillation frequency into a semiconductor chip,specifically to a technique effective in use for a control loop of atransmission VCO in a high frequency semiconductor integrated circuitthat modulates or up-converts a transmission signal, which is used in aradio communication device such as a mobile telephone capable oftransmitting and receiving signals of plural bands.

There is a system called the GSM (Global System for MobileCommunication), which is used in the field of the radio communicationdevice (mobile communication device) such as a mobile telephone, etc.This GSM system generally employs the phase modulation system called theGMSK (Gaussian Minimum Shift Keying) that shifts the phase of a carrieraccording to transmission data.

Now, in the recent mobile telephone of the GSM system, etc., a systemcalled the EDGE (Enhanced Data Rates for GMS Evolution) has been used inpractice in addition to the GMSK modulation mode. This new modulationsystem has the 3π/8 rotating8-PSK (phase shift keying) modulation modethat modulates the phase component and the amplitude component of acarrier. In contrast to the GMSK system that transmits information ofone bit per one symbol, the 3π/8 rotating8-PSK (hereunder, called 8-PSK)modulation system is able to send information of 3 bits per one symbol.Thus, the EDGE mode is possible of a communication at a highertransmission rate (384K bps) in comparison to the GMSK mode.

As a means of realizing the modulation system that makes the phasecomponent and the amplitude component of a transmission signal eachcarry information, there is a system called the polar loop, which isconventionally known. This system separates a transmission signal intothe phase component and the amplitude component, and then appliesfeedbacks to the phase component and the amplitude component with aphase control loop and an amplitude control loop, respectively.Thereafter, an amplifier outputs to synthesize the above controlledsignals (for example, Page 162, “High Linearity RF Amplifier Design” byKenington, Peter B, Published by ARTECH HOUSE, Inc. 1979).

In recent years, in order to downsize the device and reduce the cost bydecreasing the number of components in the radio communication system,many efforts and struggles have been made to incorporate as manycircuits as possible into one or few semiconductor integrated circuits.As an example, there is a trial to incorporate a transmission oscillatorinto the semiconductor integrated circuit that possesses themodulation/demodulation function (hereunder, called RF IC). In regard tothe RF IC for making up a communication system of the GSM system, thepresent applicant, etc. have developed and proposed the one that mountsa transmission oscillator on a semiconductor chip (Patent Document 1).

[Patent Document 1]

Ser. No. 10/373046, date of application to the US patent office: Feb.26, 2003

SUMMARY OF THE INVENTION

The inventors examined a technique that incorporates a transmissionoscillator into an RF IC configuring the communication system of theEDGE system. As the result, the examination discovered the followingproblems. Here, the inventors examined the polar loop system used in theEDGE system, and the system detects, with regard to the phase controlloop, an output of the transmission oscillator or an output of the RFpower amplification circuit (hereunder, called power amplifier), andfeeds back the output to a phase comparator that compares the outputwith a reference signal. And, with regard to the amplitude control loop,the system detects the output of the power amplifier, and feeds back theoutput to an amplitude comparator that compares the output with areference signal. The present applicant, etc. have disclosed the abovepolar loop system in the patent application (Ser. No. 10/373031, date ofapplication to the US patent office: Feb. 26, 2003).

Now, there has been the mobile telephone of the dual band system capableof handling the signals of two frequency bands, such as the 880˜915 MHzband of the GSM and the 1710˜1785 MHz band of the DCS (Digital CellarSystem). And, there are some mobile telephones of the above dual bandsystem that meet the two systems, by preparing two transmissionoscillators (hereunder, called transmission VCOs) corresponding to therespective frequency bands, and switching the transmission VCOs.

In recent years, however, the market demands the mobile telephone of atriple band system, for example, capable of handling the signal of the1850˜1915 MHz band of the PCS (Personal Communication System) inaddition to the GSM and DCS. And the market demand is estimated toincrease toward the mobile telephone that meets still more systems. Thetransmission VCO used in the mobile telephone that meets plural systemsis required to have a wide oscillation band (frequency range possible ofoscillation).

On the other hand, the Q-factor of the passive elements on thesemiconductor chip, such as inductors and so forth, plays a significantrole in governing the performance of the transmission VCO built in thesemiconductor chip. The Q-factor of the passive elements formed on thesemiconductor chip becomes necessarily lower than the Q-factor of thediscrete components, which is inevitable with the present semiconductormanufacturing technique. And some of them use externally mountedelements. But this cannot achieve a sufficient reduction of the numberof components. Therefore, to mount the elements including the inductorson one chip is imperative.

However, since the Q-factor of the passive elements mounted on a chip islower, the oscillation band of the VCO becomes narrower than theoscillation band of an oscillation module made up with discretecomponents, which is disadvantageous. In order to solve such a problem,the priorly filed invention (Ser. No. 10/373046, date of application tothe US patent office: Feb. 26, 2003) adopts the transmission VCO of themulti-band system. This system provides such a configuration thatgradually varies the capacitances of the transmission VCO, and selectsany of the capacitances according to the usable frequency band tothereby switch the oscillation frequencies.

However, the priorly filed invention is confined to the technique forincorporating a transmission VCO into an RF IC for the GSM system thatsupports only the GMSK modulation. The inventors examined the technicalproblems appearing in incorporating the transmission VCO of themulti-band system into the RF IC for the EDGE system. As the result, theexamination discovered that, as the capacitances of the on-chipcapacitors constituting the transmission VCO deviate from the designvalue due to the manufacturing dispersions, the KV characteristic(voltage vs. frequency sensitivity) of each band deviates from thedesired characteristic, which varies the loop gain of the control loop.Thereby, the EVM (Error Vector Magnitude) deteriorates, and also thespectral regrowth (transmission spectrum) inevitably deteriorates, whichsignifies the attenuation of a signal level at the frequency deviatingby 0.4 MHz from the carrier frequency.

Now, as the EVM deteriorates, the modulation accuracy decreases; and asthe spectral regrowth deteriorates, the leakage of noises into theadjacent channels increases. Therefore, it is necessary to prevent boththe deterioration of the EVM and the deterioration of the transmissionspectrum as much as possible. Here, in the polar loop system thatexecutes the phase control and the amplitude control separately, thesignal of the transmission VCO only contains the phase modulationcomponent, and does not contain the amplitude modulation component. Butnevertheless, the EVM deteriorates due to the dispersion of the KVcharacteristic of the transmission VCO. The reason is considered suchthat there is a close relation between the phase and the amplitude inthe PSK modulation, and the error of the phase affects the error of theamplitude.

It is therefore an object of the invention to provide, when atransmission oscillator is incorporated into a semiconductor chip, acommunication semiconductor integrated circuit (RF IC) capable ofsecuring the oscillation operation over a wide frequency range, andenhancing the accuracy of the oscillation frequency.

Another object of the invention is to provide, when a transmissionoscillator is incorporated into a semiconductor chip, a communicationsemiconductor integrated circuit (RF IC) capable of preventing thedeterioration of the EVM and the deterioration of the transmissionspectrum, due to the variation of the KV characteristic of thetransmission oscillator accompanied with the manufacturing dispersions.

Another object of the invention is to provide a technique effective inuse for incorporating a transmission oscillator into a communicationsemiconductor integrated circuit that configures the EDGE system toexecute the phase modulation and the amplitude modulation.

The foregoing and other objects and the novel features of the inventionwill become apparent from the descriptions and appended drawings of thisspecification.

An outline of representative ones of inventions disclosed in thespecification will be briefly described as follows.

According to one aspect of the invention, in the phase control loopincluding the transmission oscillator (transmission VCO), thecommunication semiconductor integrated circuit corrects a dispersion ofthe KV characteristic of the transmission VCO by calibrating a currentIcp of a charge pump inside the phase control loop. More in concrete,the integrated circuit measures a KV value Kv (oscillation frequencyrange/control voltage range) of the transmission VCO, and calibrates thecurrent Icp of the charge pump so that Kv Icp falls into a predeterminedvalue. And, in correcting the dispersion of the KV characteristic of thetransmission VCO by the current Icp of the charge pump, if the currentIcp of the charge pump has a deviation, a correct calibration will notbe achieved. Therefore, the integrated circuit first measures thecurrent Icp of the charge pump and corrects the deviation of the currentIcp, prior to correcting the KV characteristic of the transmission VCO.

According to the above means, in the phase control loop including thetransmission VCO, when the KV characteristic of the transmission VCOdisperses, the current Icp of the charge pump is calibrated so thatKv·Icp falls into a constant value, which makes it possible to preventthe deterioration of the EVM and enhance the modulation accuracy.

Especially, the invention is effective in use for incorporating atransmission oscillator into the communication semiconductor integratedcircuit that configures the EDGE system to execute the phase control andamplitude control of the 8-PSK-modulation system by the polar loopsystem. The reason is as follows.

In the polar loop system, if the KV characteristic of the transmissionVCO deviates, the loop gain of the phase control loop will vary;accordingly, there is the apprehension that the transmission spectrumdeteriorates. Concretely, the frequency characteristic of the loop gainof the phase control loop having two poles ωp1, ωp2 and one zero pointωz is shown in FIG. 12. Since the positions of the poles and the zeropoint are fixed by the loop filter, if the KV characteristic of thetransmission VCO deviates, the frequency characteristic of the loop gainwill shift up and down as shown by the dotted lines in the drawing.Therefore, the noise suppression of the phase control loop fluctuatesdue to the dispersion of the KV characteristic, which invitesdeterioration of the transmission spectrum. Therefore, the above meansis able to enhance the transmission spectrum by correcting thedispersion of the KV characteristic of the transmission VCO.

And in the polar loop system, the transmission signal is divided intothe phase component and the amplitude component, which are controlled bythe phase control loop and the amplitude control loop, respectively;thereafter, they are synthesized and outputted by the power amplifier.Therefore, if the frequency band of the phase component is notcoincident with the frequency band of the amplitude component, themodulation accuracy will deteriorate. Table 1 shows a simulation resultof the modulation accuracy (EVM), in which the transmission circuitadopting the polar loop system is operated by the 8-PSK modulation mode,while the frequency bands of the phase control loop and the amplitudecontrol loop are varied in many ways.

TABLE 1 PM Loop Open Loop Bandwidth (MHz) 0.647 1.151 1.8 2.047 3.6396.467 AM Loop Open Loop 0.647 5.99 4.84 Bandwidth (MHz) 1.151 2.03 1.381.30 1.29 1.8 2.047 1.64 0.65 0.45 0.42 3.639 1.60 0.51 0.20 0.14 6.4673.69 1.59 0.51 0.16 0.06 (%)

Table 2 shows the spectral regrowth attained by the simulation of thesimilar condition.

TABLE 2 PM Loop Open Loop Bandwidth (MHz) 0.647 1.151 1.8 2.047 3.6396.467 AM Loop Open Loop 0.647 −46 −42 Bandwidth (MHz) 1.151 −50 −54 −52−51 1.8 2.047 −47 −60 −61 −59 3.639 −46 −58 −69 −68 6.467 −41 −46 −57−69 −73 (dBm/100 kHz)

In Table 1 and Table 2, the values on the diagonal line from the upperleft toward the lower right are those to show the state that thefrequency bands of the two control loops are coincident. Table 1 andTable 2 confirm that, as the frequency bands of the phase control loopand the amplitude control loop are more approximate to coincidence, themodulation accuracy and the spectral regrowth become better. Therefore,it is preferred that the frequency band of the phase control loop andthe frequency band of the amplitude control loop are set to an identicalband (for example, carrier frequency ±1.8 MHz) in each of the loopfilters.

Although the frequency bands are set in such a manner, if the KVcharacteristic of the transmission VCO disperses and the relationbetween the oscillation frequency range of the VCO and the loop band ofthe phase control loop deviates from the desired relation, the relationbetween the oscillation frequency range of the VCO and the loop band ofthe amplitude control loop will deviate from the desired relation.Therefore, there is the apprehension that the phase control accuracy andthe amplitude control accuracy deteriorate. However, by correcting theKv·Icp of the VCO, the relation between the oscillation frequency rangeof the VCO and the loop band of the phase control loop and the relationbetween the oscillation frequency range of the VCO and the loop band ofthe amplitude control loop can be made to fall into the desiredrelation.

Thereby, it is possible to enhance not only the phase control accuracybut also the amplitude control accuracy.

Effects obtained by representative inventions disclosed in thespecification will be briefly described as follows.

According to the invention, since the transmission oscillator (TXVCO) isconfigured in a multi-band, when the TXVCO is incorporated into asemiconductor chip, the TXVCO achieves an oscillation operation over awide frequency range. And if the KV value deviates due to thecapacitance dispersions of the capacitors, the deviation can becorrected, and the accuracy of the oscillation frequency can beenhanced.

Further, in case the TXVCO is incorporated into a communicationsemiconductor integrated circuit (RF IC) that configures the EDGE systemto execute the modulation of the phase component and the modulation ofthe amplitude component, if the KV characteristic of the TXVCO deviatesfrom the desired characteristic due to the manufacturing dispersions,the EVM will deteriorate and the transmission spectrum will alsodeteriorate. However, applying the above embodiment will correct thedeviation of the KV value of the TXVCO. Therefore, it is possible toimprove the EVM and enhance the modulation accuracy, and to prevent thetransmission spectrum from deteriorating.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a circuit configuration of a communication semiconductorintegrated circuit (RF IC) of the multi-band system relating to oneembodiment of the present invention, and it also shows a configurationexample of a radio communication system using the RF IC;

FIG. 2 is a block diagram showing one embodiment of a charge pump on aPLL loop including the TXVCO contained in the communicationsemiconductor integrated circuit (RF IC) shown in FIG. 1, and a currentmeasurement correction circuit that measures a current of a currentsource of the charge pump and generates a current correction value;

FIG. 3 is a timing chart showing the operation timing of the currentmeasurement correction circuit shown in FIG. 2;

FIG. 4 is a circuit diagram showing a concrete example of the TXVCO;

FIG. 5 is a characteristic chart showing the relation between thecontrol voltage Vt and the oscillation frequency fTX in the TXVCO;

FIG. 6 is a characteristic chart in which a part of the control voltagevs. oscillation frequency characteristic in the TXVCO is expanded;

FIG. 7 is a characteristic chart showing the relation between theoscillation frequency difference and the KV variation at two points inthe same band, in the TXVCO for the GSM;

FIG. 8 is a characteristic chart showing the relation between theoscillation frequency difference and the KV variation at two points inthe same band, in the TXVCO for the DCS/PCS;

FIG. 9 is a characteristic chart showing the relation of the KVcharacteristic of each band #0 ˜#15 of the TXVCO;

FIG. 10 is a block diagram showing one embodiment of a PLL circuitincluding the IFVCO contained in the communication semiconductorintegrated circuit (RF IC) shown in FIG. 1, and a PLL circuit includingthe TXVCO contained in the same;

FIG. 11 is a timing chart showing the timing of the determination of ausable band and the correction of a current value, based on thefrequency measurements of each VCO and the measured results thereof, inthe radio communication system using the RF IC relating to the oneembodiment of the invention; and

FIG. 12 is a loop gain frequency characteristic chart of the phasecontrol loop in the polar loop system.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Next, the embodiment of the invention will be described with theaccompanying drawings.

FIG. 1 is a block diagram that shows one embodiment of a communicationsemiconductor integrated circuit (RF IC) incorporating a transmissionoscillator relating to the invention, and a configuration example of aradio communication system using the RF IC. This embodiment was appliedto a radio communication system of the EDGE system that executes the8-PSK modulation by the so-called polar loop system having the phasecontrol loop and the amplitude control loop.

The radio communication system shown in FIG. 1 includes an antenna 100that transmits/receives radio wave signals, a switch 110 that switchesthe transmission/reception, a band pass filter 120 that removesundesired signals from a reception signal, being made up with a SAWfilter and so forth, an RF power amplification circuit (power amplifier)130 that amplifies a transmission signal, an RF IC 200 that demodulatesthe reception signal and modulates the transmission signal, and a baseband circuit 300 that converts transmission data into I, Q signal, andcontrols the RF IC 200. In this embodiment, the RF IC 200 and the baseband circuit 300 are each formed on separate semiconductor chips, asrespective semiconductor integrated circuits.

The RF IC 200 of this embodiment is configured to be capable ofmodulating/demodulating the signal according to the four communicationsystems, GSM 850, GSM 900, DCS 1800, and PCS 1900, which is notspecifically restricted. And to correspond with the above configuration,the band pass filter 120 is composed of a filter that the receptionsignal within the frequency band of the GSM system is passed through, afilter that the reception signal within the frequency band of the DCS1800 is passed through, and a filter that the reception signal withinthe frequency band of the PCS 1900 is is passed through.

The RF IC 200 of this embodiment is broadly divided into the receptionsystem circuit, the transmission system circuit, and the control systemcircuit composed of the circuits common to the transmission/receptionsystem, such as control circuits and clock circuits other than theformer two.

The reception system circuit includes a low noise amplifier 210 thatamplifies the reception signal, a frequency divider/phase shifter 211that divides the frequency of an oscillation signal φRF generated by anRF oscillator (RFVCO) 250, and generates orthogonal signals having 90°phase shift to each other, a demodulation circuits 212 a, 212 bconstituting a mixer that demodulates the reception signal by mixing thereception signal amplified by the low noise amplifier 210 with theorthogonal signals generated by the frequency divider/phase shifter 211,high gain amplifiers (PGA) 220A, 220B that amplify the demodulated I, Qsignals, respectively, and output them to the base band circuit 300.

The transmission system circuit TXC includes an oscillator (IFVCO) 230that generates an oscillation frequency φIF of the intermediatefrequency such as 640 MHz, a frequency divider/phase shifter 232 thatdivides the frequency of the oscillation frequency φIF generated by theoscillator (IFVCO) 230, and generates orthogonal signals having 90°phase shift to each other, a modulation circuits 233 a, 233 bconstituting a mixer that modulates the generated orthogonal signals bythe I signal and Q signal supplied from the base band circuit 300, anadder 234 that mixes the modulated signals, transmission oscillators(TXVCO) 240 a, 240 b that generate a transmission signal φTX of apredetermined frequency, an offset mixer 253 a that mixes thetransmission signal φTX (feedback signal) outputted from thetransmission oscillators 240 a, 240 b with a signal φRF′ acquired bydividing the frequency of the oscillation signal φRF generated by the RFoscillator (RFVCO) 250, and thereby generates a signal of a frequencyequivalent to the frequency difference of the two signals (φTX, φRF′), aphase comparator 236 that compares an output from the offset mixer 253 aand a signal TXIF mixed by the adder 234 to detect the frequencydifference and the phase difference between the two signals, a chargepump & loop filter 237 that generates a voltage corresponding to theoutput from the phase comparator 236, a second offset mixer 235 b thatmixes a feedback signal acquired by extracting a transmission outputfrom the RF power amplifier 130 by means of a coupler or the like withthe signal φRF′ generated by the RF oscillator (RFVCO) 250, and therebygenerates a signal of a frequency equivalent to the frequency differenceof these signals, an amplitude comparator 238 that compares an output ofthe second offset mixer 235 b and the signal TXIF mixed by the adder 234to detect an amplitude difference thereof, and a PA output controlcircuit 239 that generates a signal Vapc for controlling the gain of theRF power amplifier 130 on the basis of a voltage corresponding to thedetected amplitude difference and an output level indicating signalVramp from the base band IC 300 and so forth.

One of the transmission oscillators 240 a, 240 b generates the signal ofthe 850˜900 MHz band for the GSM, and the other one generates the signalof the 1800˜1900 MHz band for the DCS and PCS.

Further, the RF IC 200 of this embodiment includes on the chip thereof acontrol circuit 260 that controls the whole chip, an RF synthesizer 261that constitutes an RF PLL circuit together with the RF oscillator(RFVCO) 250, an IF synthesizer 262 that constitutes an IF PLL circuittogether with the oscillator (IFVCO) 230 for the intermediate frequency,and a reference oscillator (VCXO) 264 that generates a clock signal φrefas the reference clock for these synthesizers 261, 262. The synthesizers261, 262 are each configured with phase comparators and charge pumps andloop filters and so forth.

Since the reference oscillation signal φref is required to have highfrequency accuracy, the reference oscillator (VCXO) 264 has a quartzresonator Xtal externally connected. With regard to the frequency of thereference oscillation signal φref, 26 MHz or 13 MHz is selected. Thequartz resonator of such a frequency is a general-purpose component, andis readily available.

The control circuit 260 contains a control register, and the setting isexecuted to this control register on the basis of a signal from the baseband IC 300. Concretely, the base band IC 300 supplies the RF IC 200with a clock signal CLK for synchronization, a data signal SDATA, and aload enable signal LEN as the control signal. As the load enable signalLEN is asserted to the effective level, the control circuit 260sequentially fetches the data signal SDATA being transmitted from thebase band IC 300 synchronously with the clock signal CLK, and sets thesignal to the control register. The data signal SDATA is transmitted inserial order, which is not specifically restricted. The base band IC iscomposed of a microprocessor and so forth.

The control register contained in the control circuit 260 includes thecontrol bits for starting the frequency measurements of the VCOs in theRF oscillator (RFVCO) 250 and the intermediate frequency oscillator(IFVCO) 230, and the bit fields for designating the mode such asreception mode, transmission mode, idle mode, etc. Here, the idle modesignifies the sleep state that only extremely limited circuits are inoperation during the wait state, and major part of the circuitsincluding at least the oscillators are in halt.

In this embodiment, the phase comparator 236, charge pump & loop filter237, transmission oscillators (TXVCO) 240 a, 240 b, and offset mixer 236constitute a transmission PLL circuit (TXPLL) that executes thefrequency conversion. In the radio communication system of themulti-band system in this embodiment, on the basis of the command fromthe base band IC 300, the control circuit 260 varies the frequency ofthe oscillation signal φRF of the RF oscillator (RFVCO) 250 incorrespondence with the channel used, during the transmission andreception; and it also varies the frequency of the signal supplied tothe offset mixers 235 a, 235 b in correspondence with the GSM mode orthe DCS/PCS mode. Thereby, the switching of the transmission frequencyis carried out.

On the other hand, the oscillation frequency of the RF oscillator(RFVCO) 250 is set to a different value in correspondence with thereception mode or the transmission mode. In the transmission mode, theoscillation frequency fRF of the RF oscillator (RFVCO) 250 is set to3616˜3716 MHz in case of the GSM 850, 3840˜3980 MHz in case of the GSM900, 3610˜3730 MHz in case of the DCS, and 3860˜3980 MHz in case of thePCS. The oscillation frequency fRF is divided into ¼ by the frequencydivider in case of the GSM, and it is divided into ½ in case of the DCSand PCS, which is supplied to the offset mixers 235 a, 235 b.

The offset mixer 235 a outputs a signal equivalent to a frequencydifference (fRF−fTX) of the oscillation signal φRF from the RFoscillator (RFVCO) 250 and the transmission signal φTX from thetransmission oscillators (TXVCO) 240 a, 240 b. In order that thefrequency of the signal equivalent to the frequency difference becomesequal to the frequency of the modulation signal TXIF, the transmissionPLL circuit (TXPLL) operates. In other words, the transmissionoscillators (TXVCO) 240 a, 240 b are controlled to oscillate at afrequency equivalent to the difference between the frequency (fRF/4 incase of the GSM, fRF/2 in case of the DCS and PCS) of the oscillationsignal φRF from the RF oscillator (RFVCO) 250 and the frequency of themodulation signal TXIF.

And in this embodiment, in order to calibrate the frequency of thetransmission oscillators (TXVCO) 240 a, 240 b and to select the usableband, the signal φTX outputted from the transmission oscillators (TXVCO)240 a, 240 b is supplied to the IF synthesizer 262, and band selectionsignals VB3˜VB0 from the IF synthesizer 262 are supplied to thetransmission oscillators (TXVCO) 240 a, 240 b.

In the RF IC of this embodiment, the KV value Kv (=oscillation frequencyrange/control voltage range) of the transmission VCO is measured, thecurrent Icp of the charge pump is calibrated so that Kv·Icp falls into apredetermined value, and thereby the dispersion of the KV characteristicin the transmission VCO is corrected. When the dispersion of the KVcharacteristic in the transmission VCO is corrected by means of thecurrent Icp of the charge pump, if the current Icp of the charge pumpdeviates, any correct calibration will not be possible. Therefore,before the correction of the KV characteristic in the transmission VCO,it is essential to measure the current of the charge pump first, and tocorrect the deviation.

The method of correcting the deviation of the current Icp of the chargepump of this embodiment will be described with FIG. 2. FIG. 2 shows thedetail of a part of the charge pump 237 a of the charge pump & loopfilter 237 in FIG. 1.

The charge pump 237 a includes a current source IP for charging up, acurrent source IN for charging down, switches S1, S2 connected betweenthese current sources IP, IN and an output node N0, which are turned onor off by the up signal UP or the down signal DOWN from the phasecomparator 236 during transmission, current sources T11˜T1n for thecurrent correction, which are placed in parallel to the current sourceIP, and calibration switches S11˜S1n provided between these currentsources T11˜T1n and the switch S1 and so forth.

A capacitor CLF being a constituent of the loop filter is connectedbetween the output node N0 of the charge pump 237 a and the ground. Thecapacitor CLF is an externally mounted capacitor in this embodiment,which is not specifically restricted. When the loop filter is of thefirst order, it can be configured with the capacitor CLF and a wiringparasitic resistor and so forth. When the loop filter is of a higherorder, higher than the second order, an additional capacitor andresistor have to be prepared. The capacitor and resistor except for thecapacitor CLF may be placed before the phase comparator 236.

The charge pump 237 of this embodiment accompanies a current measurementcorrection circuit 370 capable of measuring the current value of thecurrent source IP and correcting the current dispersion. The currentmeasurement correction circuit 370 includes a comparator 371 thatdetects the potential at the output node N0 of the charge pump 237 a, aD-type flip flop 372 that operates as a latch by the output of thecomparator 371, a timer counter 373 that counts the reference clock φrefto perform a timer operation, an arithmetic circuit 374 that calculatesa correction value of the current Icp based on a counter value of thetimer counter 373 and a defined value, and a register 375 that storesthe correction value calculated and so forth. And the arithmetic circuit374 supplies signals TD1˜TDn for controlling the calibration switchesS11˜S1n inside the charge pump 237 a, thereby correcting the deviationof the current value of the current source IP. Further, a reset switchS0 for discharging the charges of the capacitor CLF is provided betweenthe output node N0 of the charge pump 237 a and the ground.

Next, the operation of measuring and correcting the current value of thecurrent source IP of the charge pump 237 with be described with FIG. 3.

In measuring the current value, first, a measurement starting signalTX_ON is set to Low level, the switch S0 is turned on, the charges ofthe capacitor CLF are discharged, and in this condition, a signalICPDEF_ON is brought to High level, thus activating the comparator 371(timing t1). And the switch S1 is set to the on state, and the switch S2and switches S11˜S1n are set to the off state. And after 1 μsec, forexample, the measurement starting signal TX_ON is turned to High level(timing t2). Then, the switch S0 is turned off, the capacitor CLF ischarged by the current from the current source IP, and the potentialVcap at the output node N0 is increased gradually. Accompanied withthis, the timer counter 373 starts the timer operation based on theclock φref.

When the current value of the current source IP is the value as set, thevoltage that the potential Vcap at the output node N0 is to reach at apredetermined time after starting the charge is applied to thecomparator 371 as a comparison voltage VBG. As the potential Vcap at theoutput node N0 reaches the comparison voltage VBG, the output of thecomparator 371 varies to change the output of the flip flop 372 intoHigh level (timing t3). Thereby, the timer counter 373 halts the timeroperation. The arithmetic circuit 374 fetches the coefficient of thetimer counter at that moment, compares it with the defined valueprovided in advance, thereby detects how far the current Icap of thecurrent source IP deviates from the design value, and generates thecontrol signals TD1˜TDn according to the deviation detected.

Here, when the current value of the current source IP is the value asdesigned, it is given by Icp, and when the current value of the currentsource IP deviates from the design value, it is given by Icp′; and whenthe capacitor CLF is charged by Icp and the voltage Vcap reaches thecomparison voltage VBG, the counter value of the counter 373 is given byCNT, and when the capacitor CLF is charged by Icp′ and the voltage Vcapreaches the comparison voltage VBG, the counter value of the counter 373is given by CNT′. On the above definitions, the ratio CNT′/CNT isassumed as the current correction coefficient A. Then in thisembodiment, the arithmetic circuit 374 and the current sources T11˜T1nfor the current correction are configured to be capable of correctingthe current Icp by 1% when A is ‘+1’, and correcting the current Icp by2% when A is ‘+2’, . . . . Thereby, the control signals TD1˜TDn to theswitches S11˜S1n inside the charge pump 237 can easily be generated fromthe current correction coefficient A.

The control signals TD1˜TDn determine the on/off state of the switchesS11˜S1n, which determines the current sources T11˜T1n for the currentcorrection that are connected in parallel with the current source IP.The current sources T11˜T1n for the current correction have smallervalues than the current source IP, and use the values weighted by then-th power of 2. Thereby, comparably few current sources perform thecurrent correction with comparably large number of stages.

FIG. 4 shows a circuit configuration of the transmission VCO used inthis embodiment. The transmission VCO used in this embodiment is an LCresonant oscillator, and it includes a pair of N-channel MOS transistorsQ1, Q2 whose sources are commonly connected, and whose gates and drainsare cross connected, a constant current source I0 connected between thecommonly connected sources of the transistors Q1, Q2 and the ground GND,inductors L1, L2 connected between each of the drains of the transistorsQ1, Q2 and a power supply terminal Vcc, a capacitor C1, varactor diodesDv1, Dv2 as a variable capacitor, and a capacitor C2 that are connectedin series between the drain terminals of the transistors Q1, Q2, aninductor L11 connected between a connection node n1 of the capacitor C1and the varactor diode Dv1 and the ground, an inductor L12 connectedbetween a connection node n2 of the capacitor C2 and the varactor diodeDv2 and the ground, capacitors C11, C12 connected in series between thedrain terminals of the transistors Q1, Q2, and capacitors C21, C22; C31,C32; C41, C42 that are connected in parallel to the capacitors C11, C12,and so forth.

And in the oscillator of this embodiment, the control voltage Vt fromthe charge pump & loop filter 237 is applied to a connection node n0 ofthe varactor diode Dv1 and the varactor diode Dv2, and the oscillatorfrequency is controlled to be continuously variable. On the other hand,the band selection signals VB3˜VB0 are supplied from a conformable banddetermination circuit 19 to a connection node n11 of the capacitor C11and the capacitor C12, a connection node n12 of the capacitor C21 andthe capacitor C22, a connection node n13 of the capacitor C31 and thecapacitor C32, and a connection node n14 of the capacitor C41 and thecapacitor C42. Thus, the band selection signals VB3˜VB0 are brought toeither High level or Low level, thereby the oscillator frequency iscontrolled to be stepwise variable.

The capacitor C11 and the capacitor C12 have a same capacitance, and inthe same manner, C21 and C22, C31 and C32, C41 and C42 each have a samecapacitance. The capacitances of the capacitors C11, C21, C31, and C41are set to each have the weighting of the m-th power of 2 (m: 3, 2, 1,0), and the capacitances can be varied with 16 steps corresponding tothe combinations of the band selection signals VB3˜VB0. Therefore, theoscillator is controlled to operate on any of the frequencycharacteristics of 16 bands as shown in FIG. 5.

In a trial of widening the frequency range that the VCO has to cover, ifthe trial is made only with the capacitance variations of the varactordiodes Dv1, Dv2 by the control voltage Vt, the Vt-frF characteristicwill become sharp, as shown by the dotted line A in FIG. 5, and thesensitivity of the VCO, namely, the ratio of the frequency variationagainst the control voltage variation (Δf/ΔVt) will be increased.Consequently, the VCO becomes susceptible to noises. That is, only aslight noise superposed on the control voltage Vc will greatly vary theoscillation frequency of the VCO.

To solve this problem, the transmission oscillators (TXVCO) 240 a, 240 bof this embodiment contain plural capacitors connected in parallel thatconstitute the LC resonant oscillator, switch the capacitors to beconnected into 16 steps by the band selection signals VB3˜VB0, and varythe capacitance C. Thus, the transmission oscillators (TXVCO) 240 a, 240b are configured to perform the oscillation control according to the 16Vt-ftF characteristic curves as shown by the solid lines in FIG. 5, andto select any one of the characteristics in correspondence with theusable frequency band during transmission.

Here, in the LC resonant oscillator of this embodiment, the capacitorsC11˜C42 are formed with capacitors between the gate electrodes ofN-channel MOS transistors and the substrate. And, by appropriatelysetting the ratio of the gate widths of the MOS transistors forming thecapacitors C11˜C42, a desired capacitance ratio (m-th power of 2) can beattained. Hereunder, the capacitors C11˜C42 are called band switchingcapacitors, and the varactor diodes Dv1 and Dv2 are called variablecapacitors. Capacitors formed on a semiconductor substrate, having asandwiched structure of metal film-insulating film-metal film, may beused for the capacitors C11˜C42.

In this embodiment, on-chip devices are used for the inductors L1, L2,L11, and L12. This is to reduce the number of components, and naturallyexternal elements can be used. The reason for adding the inductors L11,L12 other than the inductors L1, L2 is to reduce the dependence of theoscillation frequency upon the power supply voltage Vcc, and theinductors L11, L12 and the capacitors C1, C2 can be eliminated. In thatcase, the connection of the varactor diodes is inverted.

Next, the measurement of the frequencies of respective bands and thecorrection operation of the KV characteristic in the transmission VCOwill be explained. The dispersion of the KV characteristic in thetransmission VCO largely depends on the capacitance dispersions of thevariable capacitors Dv1, Dv2 and the capacitance dispersions of the bandswitching capacitors C11˜C42. First, the relation between thecapacitance dispersions of the variable capacitors Dv1, Dv2 and thedispersion of the KV characteristic will be explained with FIG. 6through FIG. 8.

The frequency dispersion of the VCO dependent on the capacitancedispersions of the variable capacitors Dv1, Dv2 appears at the maximum,when the High level voltage is applied to the nodes n11˜n14 to disappearall the band switching capacitors C11˜C42. The frequency characteristicof the VCO in this case is the band #15 out of the bands #0˜#15 as shownin FIG. 5. FIG. 6 shows the characteristic of the band #15, which isexpanded. In FIG. 6, the dotted lines show the characteristic when thereare not any capacitance dispersions in the variable capacitors Dv1, Dv2,and the solid line shows an example of the characteristic when thevariable capacitors Dv1, Dv2 deviate from the design value.

FIG. 7 shows the relation of the KV variation against the frequencydifference (f1′−f2′) of the oscillation frequencies f1′, f2′corresponding to the two control voltages Vt1 (0 V), Vt2 (Vcc) of thetransmission VCO 240 a for the GSM of the transmission oscillators(TXVCO) 240 a, 240 b, based on a simulation result. FIG. 8 shows therelation of the KV variation against the frequency difference (f1′−f2′)of the oscillation frequencies f1′, f2′ of the transmission VCO 240 bfor the DCS and PCS, based on a simulation result. Here, the KVvariation is given by Kv′/Kv, in which Kv is the KV value when there arenot any capacitance dispersions in the variable capacitors Dv1, Dv2,namely, the design value, and Kv′ is the KV value of the actual VCO.FIG. 7 and FIG. 8 confirm that the KV variation is proportional to thefrequency difference f1′−f2′ (hereunder, written as Δf′) of theoscillation frequencies f1′, f2′ in the transmission VCO as shown inFIG. 4.

In this embodiment, the measurement was made as to the oscillationfrequency f1′ of the band #15 of the VCO when 0 V is applied as thecontrol voltage Vt to the node n0 on the cathodes of the varactor diodesDv1, Dv2, and the oscillation frequency f2′ of the band #15 when 2.8 Vis applied as the control voltage Vt. And, the determination was made asto a variation of the KV value from the difference Δf′ of the twofrequencies, and further a capacitance correction coefficient B forcorrecting the current value Icp′ of the current source IP of the chargepump, which is necessary for correcting the above variation of the KVvalue.

In concrete, the current value Icp′ of the current source IP of thecharge pump is calibrated to satisfy Kv·Icp=Kv′·Icp′. From thisequation, Icp′=Icp·(Kv/Kv′) is calculated. As mentioned above, thefrequency difference Δf′ is proportional to the KV variation (Kv′/Kv),namely, □f′ is unproportional to Kv/Kv′. Therefore, the above equationis expressed by Icp′=Icp·{B/Δf′}, using the frequency difference Δf′.

This confirms that if the frequencies f1′, f2′ of the transmissionoscillator (TXVCO) 240 when the control voltage Vt is set to Vt1 (0 V)and Vt2 (Vcc) are detected by measurement, Icp′ can be calibrated tosatisfy Kv·Icp=Kv′·Icp′. Besides, in the RF IC 200 of this embodiment,the current of the charge pump is controlled so as to be capable ofcorrecting the current Icp by 1% when the capacitance correctioncoefficient B is ‘+1’, and correcting the current Icp by 2% when thecapacitance correction coefficient B is ‘+2’, . . . . Thereby, thecontrol signals TD1˜TDn to the switches S11˜S1n inside the charge pump237 can easily be generated from the capacitance correction coefficientB. Here, in this embodiment, Vcc is specified as 2.8 V, but it is notlimited to this.

Further, the RF IC 200 of this embodiment provides the PLL circuit ofthe oscillator IFVCO 230 for determining the usable band in the IFVCOconfigured in multi-bands with a function to measure the frequency,which is the same as in the TXVCO. The frequencies f1′, f2′ of the TXVCOare measured by means of this function, and the usable band in the TXVCOis determined. Also, the deviation of the KV characteristic of the TXVCOaccompanied with the capacitance dispersions of the varactor diodes Dv1,Dv2 can be corrected by means of the measured values attained by thatfrequency measurement. The measurement of the frequency in the IFVCO andTXVCO will be explained afresh later.

The above KV correction is related to the KV characteristic of thehighest frequency band #15 among the KV characteristics of 16 bands thatthe TXVCO possesses, as shown in FIG. 5. The KV characteristics of theother bands #0˜#14 are determined by the connection conditions of theband switching capacitors C11˜C42, as already mentioned. And if thereare any dispersions in the band switching capacitors C11˜C42, it willproduce a deviation on the control voltage vs. oscillation frequencycharacteristic. Next, the method of correcting the dispersion of the KVcharacteristic due to the capacitance dispersions of the band switchingcapacitors C11˜C42 inside the TXVCO will be explained. Since the bandswitching capacitors C11˜C42 are formed on one and the samesemiconductor chip, if any one capacitance disperses, the othercapacitances will disperse at the same rate.

FIG. 9 shows the KV characteristic curves of each band #0˜#15. And FIG.9A shows the KV characteristic curves when the band switching capacitorsC11˜C42 are formed as the capacitances thereof are equal to the designvalues, and FIG. 9B shows the KV characteristic curves when thecapacitances of the band switching capacitors C11˜C42 deviate towardsmaller values from the design values. In this embodiment, when the KVvalue of the band #15 is given by Kv(#15), and the KV value of the band#0 is given by Kv(#0), the ratio Kv(#15)/Kv(#0) is designed to become1.55 in the TXVCO 240 a for the GSM, and it is designed to become 1.30in the TXVCO 240 b for the DCS and PCS, which is not specificallyrestricted.

As it is clear from the comparison of FIG. 9A and FIG. 9B, when thecapacitances of the band switching capacitors C11˜C42 deviate towardsmaller values against the design values, the whole distribution of thecharacteristic curves is proportionally shrunk to the direction that theratio Kv(#15)/Kv(#0) becomes smaller than 1.55 or 1.3. Although notillustrated, when the capacitances of the band switching capacitorsC11˜C42 deviate toward larger values against the design values, thewhole distribution of the characteristic curves is proportionallyexpanded to the direction that the ratio Kv(#15)/Kv(#0) becomes largerthan 1.55 or 1.3.

Here, the KV value Kv(#15)′ of the band #15 in FIG. 9B (in the state ofall the band switching capacitors C11˜C42 being off) deviates from theKV value Kv(#15) of the band #15 in FIG. 9A. This comes from that the KVcharacteristic curves in FIG. 9 were shown to include the capacitancedispersions of the variable capacitors Dv1, Dv2. If the deviation of theKV value accompanied with the capacitance dispersions of the variablecapacitors Dv1, Dv2 is corrected by the above correction method, the KVvalue Kv(#15)′ of the band #15 will coincide with the Kv(#15) of thedesign value. Therefore, the KV characteristic distribution of the TXVCOafter the above correction is found to become almost proportional to theKV characteristic distribution of the idealistic TXVCO.

The bottoms of the parabolas showing the KV characteristics (the minimumvalues of the KV) of each band sit side by side on a straight line, asshown in FIG. 9. And, the KV value Kv(#15) of the band #15 of the designvalue and the KV value Kv(#0) of the band #0 can be known in advance bycalculation. Therefore, if the frequency f15′ of the band #15 and thefrequency f0′ of the band #0 are measured in the actual VCO, the KVvalue Kv(#15) of the band #15 can be obtained by calculation. And, withregard to any two bands other than the band #15 and the band #0, forexample, the band #6 and the band #7,(f0-f15):(f6-f7)=(f0′-f15′):(f6′-f7′) is satisfied.

Therefore, the deviation of the KV value can be calculated by means ofmeasuring the frequencies of arbitrary two bands other than the band #15and the band #0, whereby the correction value of the current Icp of thecurrent source of the charge pump can be determined. In the aboveexplanation, the frequencies of each band are defined as the frequenciesat the bottom positions of the KV characteristic carves; however, sincethe KV characteristic curves of each band are virtually the same, theintersections of a line B′ made by the parallel displacement of avirtual line B in FIG. 9A and the KV characteristic curves of each bandsit on the same positions of the characteristic curves. Therefore, ifthe frequencies corresponding to the intersections of the line B′ andthe KV characteristic curves of each band, namely, the control voltagesVt are the same, the correction value can be determined by using thefrequency measurement value of each band at an arbitrary control voltage(for example, 1.0 V).

Further, if the correction value is determined on the basis of thefrequencies f6′, f7′ of the TXVCO that are measured without correctingthe capacitance dispersions of the variable capacitors Dv1, Dv2, thedetermined correction value is totalized with a correction value that isdetermined to correct the KV value due to the capacitance dispersions ofthe variable capacitors Dv1, Dv2, and the totalized value is used as anew correction value,

it will be possible to correct the deviation of the KV value due to thecapacitance dispersions of the variable capacitors Dv1, Dv2, and thedeviation of the KV value due to the capacitance dispersions of the bandswitching capacitors C11˜C42 at the same time. In that case, it ispossible to store the correction value after totalized in the register37, to read out the correction value corresponding to a value by which aconformable band is determined, and to supply it to the charge pump.Concretely, it is preferred to calculate the correction valuescorresponding to the frequency measurement values of each band stored inthe register 37, and to store the values in the register 37 as atable-format data.

It is the same with the dispersion of the current Icp of the currentsource of the charge pump; and provided that the charge pump is operatedwith the current including the dispersion, and to (or from) thecorrection value of the KV value attained from the measured frequency isadded (or subtracted) a correction value that is determined throughdetecting the dispersion of the current Icp of the current source of thecharge pump, and the value after the above addition (including thesubtraction) is used as a new correction value, it is possible tocorrect the deviation of the KV value due to the dispersion of thecurrent Icp of the current source of the charge pump, the deviation ofthe KV value due to the capacitance dispersions of the variablecapacitors Dv1, Dv2, and the deviation of the KV value due to thecapacitance dispersions of the band switching capacitors C11˜C42 at thesame time. Also in this case, it is possible to store the correctionvalue after totalized in the register 37.

And, also in case of correcting the deviation of the KV value due to thecapacitance dispersions of the band switching capacitors C11˜C42 by theabovementioned calibration of the current Icp of the current source ofthe charge pump, it will be convenient if the count value ‘1’ of thecorrection value is correspondent to 1% of the current Icp. Accordingly,in determining the correction value from the frequency difference, thisembodiment intended to find out any two bands in which the frequencydifference is correspondent to 1% of the current Icp, and to use thefrequency measurement values of the two bands and determine thecorrection value of the KV value. Further, the PLL circuit of the IFVCOin the RF IC 200 is provided with the counter function that measures thefrequency in order to determine the usable band, accordingly thisembodiment intended to measure the frequency of the IFVCO and thefrequency of the TXVCO by using this counter function to determine theusable band, and to select a usable value in determining the correctionvalue of the KV value among the measured frequency values in the TXVCO.

Next, the circuit to calibrate the frequency measurement value of theTXVCO and the KV value will be explained in detail. FIG. 10 shows aconcrete example of the PLL circuit, which is provided with the functionto measure the frequency of the TXVCO and the function to select theusable band on the basis of the measurement result. Here, in FIG. 10,the same circuits and devices as those shown in FIG. 1 are given thesame symbols to eliminate repeated explanations.

In the embodiment shown in FIG. 10, a current measurement correctioncircuit 370′ is the circuit in which the arithmetic circuit 374 and theregister 375 are eliminated from the current measurement correctioncircuit 370 shown in FIG. 2. The function of the arithmetic circuit 374and the register 375 is contained in an arithmetic circuit 38 and aregister 37 for determining the usable band, described later. And inFIG. 10, the circuit equivalent to the charge pump 237 is notillustrated. This is because the output stage of the phase comparator237 can be configured as the circuit that possesses the same function asthe charge pump.

On the other hand, FIG. 10 shows the circuit configuration of a loopfilter 237 b. The resistors and capacitors configuring the loop filter237 b are connected as externally mounted elements. Further in theembodiment of FIG. 10, during the frequency measurement of the TXVCO 240a, 240 b, and the drawing-in of the PLL circuit, a predetermined dcvoltage VDC from a dc power supply 217 is supplied to the loop filter238 by a switch SW0′, instead of the voltage Vt from the charge pump.

The PLL circuit shown in FIG. 10 is a concrete example of the IFsynthesizer 262 shown in FIG. 1. In this embodiment, the PLL circuit isconfigured to be capable of measuring the frequency of the oscillator(IFVCO) 230 that generates the oscillation signal φIF of theintermediate frequency by using the IF synthesizer 262 as well as thefrequency of the transmission oscillators (TXVCO) 240 a, 240 b. Withregard to the TXVCO 240 a for the. GSM and the TXVCO 240 b for the DCSand PCS, either one of them is made into operation by the control signalfrom the control circuit 260, during the frequency measurement andtransmission.

The IF synthesizer 262 includes a frequency divider DVD1 that dividesthe frequency of the oscillation signal φIF of the IFVCO 230 into ½,aselector 31 that selects the signal divided by the frequency dividerDVD1 or an output signal from a frequency divider DVD2 that divides thefrequency of the oscillation signal φTX of the TXVCO 240 a (240 b) into½ or ¼, a variable frequency divider 32 that divides a selected signalfrequency, a fixed frequency divider 33 that divides the frequency 26MHz of the reference oscillation signal φref from the referenceoscillator (VCXO) 264 into 1 MHz, and an IF PLL circuit 30 and so forth.

The detail of the IF PLL circuit 30 is not illustrated in the drawing,but it includes a phase comparator, a charge pump, and a loop filter andso forth, in the same manner as the transmission PLL circuit. Thevariable frequency divider 32 includes a pre-scaler 321 capable of 1/16frequency division or 1/17 frequency division, an N counter 322, and anA counter 323, in which the latter two constitute a modulo-n counter.

Further, the IF synthesizer 262 includes a comparator 35 that compares avalue counted by the N counter 322 during the frequency measurement ofthe VCO and the reference data (frequency information) stored in an ROM40, a counter register 36 that holds the usable band information of theIFVCO 230 based on the comparison result of the comparator 35, aregister 37 that stores a value counted by the N counter 322 during thefrequency measurement of the IFVCO and TXVCO, the arithmetic circuit 38that calculates target oscillation frequency values TX (N, A) of theIFVCO and TXVCO on the basis of frequency set values RF/IF (N, A) of theRFVCO and IFVCO being supplied from the base band circuit, and aconformable band determination circuit 39 that compares a calculatedvalue by the arithmetic circuit 38 and a value stored in the register 37to generate codes VB2′˜VB0′, VB3˜VB0 for designating the usable band inthe IFVCO and TXVCO. Here, the conformable band determination circuit 39can be configured as a part of the control circuit 260. The arithmeticcircuit 38 can be made into a common circuit to the arithmetic circuit374 shown in FIG. 2, and the register 37 can be made into a commoncircuit to the register 375 shown in FIG. 2.

During the frequency measurement, the dc voltage VDC supplied to theloop filter 16 by the switch SW0 may be any value if it is within theeffective variable range of the control voltage Vc. In this embodiment,an intermediate voltage in the variable range of the control voltage Vt,for example, 1.0 V is selected as the dc voltage VDC. Naturally, the dcvoltage VDC is not limited to 1.0 V, and an arbitrary voltage such as1.2 V or 1.4 V can be taken, if it is within the variable range of thecontrol voltage Vt. During the frequency measurement, the dc voltage VDCis designed to take one and the same value even if the band is switched.The control circuit 260 controls the switch SW0, variable frequencydivider 32, comparator 35, register 37, arithmetic circuit 38, andconformable band determination circuit 39.

The IFVCO 230 is configured with an LC resonant oscillator similar tothe TXVCO shown in FIG. 4, and plural capacitors constituting the LCresonant oscillator are arrayed in parallel. The capacitors areselectively connected by the band switching signals VB1˜VB0 to therebyswitch the capacitance C of the LC resonant oscillator, so that theoscillation frequency can be switched into four steps. In thisembodiment, the IF synthesizer 262 contains the register 37 and theconformable band determination circuit 39, which saves the calibrationwork called the frequency adjusting being performed in the conventionalPLL circuit.

That is, in the conventional PLL circuit, while operating the VCO andmeasuring the frequency, the frequency adjusting work has been carriedout so as to make the control voltage vs. frequency characteristic(Vt-flF characteristic) have a predetermined initial value and apredetermined slope. On the other hand, in the PLL circuit of thisembodiment, the switch SW0 is switched in advance to apply apredetermined dc voltage VDC to the IFVCO 230, and the frequencies ofeach band are measured to store the data in the register 37. In theactual use, the set value corresponding to a specified band that isgiven to the N counter 322 and A counter 323 from the outside iscompared with the measurement values stored in the register 37, the onethat can cover the frequency range of the specified band is selectedamong the plural (eight) Vt-flF characteristic curves, and the switchingof the IFVCO 230 (switching of the capacitors) is carried out so as toperform the oscillation control operation according to thecharacteristic curve.

According to this method, provided that the IFVCO is designed to coverthe frequency range slightly wider than the desirable range inconsideration for the dispersions, and to overlap the frequency rangelittle by little (preferably, half-and-half) by the adjacent two of theeight-step Vt-flF characteristic curves, there is always acharacteristic curve that covers the specified frequency range.Therefore, it is only needed to select the Vt-flF characteristic curvecorresponding to the specified band, on the basis of the actualcharacteristics attained by the measurement. Thereby, the frequencyadjusting becomes unnecessary, and it is not needed to set in advancethe usable band and the switching state of the IFVCO in one-to-onecorrespondence. This is the same with the TXVCO 240 a, 240 b.

The method of dividing the frequency of the oscillation signal by themodulo-n counter composed of the pre-scaler 321, N counter 322, and Acounter 323 is already a well-known technique. The pre-scaler 321 isconfigured to be capable of dividing the frequency with two differentfrequency-dividing ratios, such as 1/16 and 1/17, and the count endingsignal of the N counter 322 switches one frequency-dividing ratio intothe other one. The N counter 322 and A counter 323 are a programmablecounter. The N counter 322 receives the integral parts of the quotientsacquired when a desired frequency (the oscillation frequency fIF of theVCO to be desirably acquired as the output) is divided by the frequencyfref′ of the reference oscillation signal φref′ and the firstfrequency-dividing ratio (17, in this embodiment), as the set value; andthe A counter 323 receives the residues (MOD) thereof as the set value.The counters terminate the counting operations when counting out the setvalues, and count again the set values.

As the pre-scaler 321 and the modulo-n counter are put in operationaccording to such a procedure, first the pre-scaler 321 divides thefrequency of the oscillation signal of the IFVCO into 1/(2·16), and theA counter 323 counts the output until the set value. When terminatingthe counting, the A counter 323 outputs the count ending signal MC, andthis signal MC switches the operation of the pre-scaler 21. Until the Acounter 323 counts the set value, the pre-scaler 321 divides thefrequency of the oscillation signal of the RFVCO 250 into 1/(2·17). Byrepeating such operations, the modulo-n counter 2 becomes able to dividethe frequency of the oscillation signal not by the integral ratio, butby the ratio having a decimal part.

Further, this embodiment is configured to switch the selector 31 andinput the oscillation signal from the TXVCO 240 a or 240 b to thepre-scaler 321; and the N counter 322 is configured to be able tooperate as an 11-bits counter during the frequency measurement of theTXVCO. Thereby, the TXVCO 240 a and 240 b are configured to be able toswitch the oscillation frequency into 16 bands, namely, 16 steps. Thefrequency is measured as to 15 bands (#0˜#14) among 16 bands. Themeasured frequencies in each band (#0˜#14) are stored in the register37. In other words, in case the TXVCO has 16 switching bands, thecounter is only needed to measure the calibration values of 15 bands.When the usable frequency band during the transmission does not conformto the bands #0˜#14, the band #15 is automatically selected.

The control circuit 260 generates, during the frequency measurement ofthe IFVCO, the switching signals VB2′˜VB0′ to sequentially select 8bands, and outputs them to the IFVCO 230. And the control circuit 260generates, during the frequency measurement of the TXVCO, the switchingsignals VB3˜VB0 to sequentially select 16 bands, and outputs them to theTXVCO 240 a, 240 b. Further, when the power supply is turned on, thecontrol circuit 260 puts the current measurement correction circuit 370in operation. And the control circuit 260 controls the circuit 370 tomeasure the current of the current source of the charge pump 237 a, todetect the dispersion of the current, to determine the currentcorrection value, and controls the register 37 to store the currentcorrection value.

During the transmission, the conformable band determination circuit 39compares the measurement values stored in the register 37 and the setcodes IF (N, A) of the N counter 322 and the A counter 323 supplied tothe arithmetic circuit 38, generates 3-bits codes VB2′˜VB0′, and outputsthem to the IFVCO 230 as the band switching signal. And the conformableband determination circuit 39 compares the value calculated by thearithmetic circuit 38 on the basis of the set codes RF (N, A) and IF (N,A) and the value stored in the register 37, generates 4-bits codesVB3˜VB0, and outputs them to the TXVCO 240 a, 240 b as the bandswitching signal. Since the RF IF 200 contains the arithmetic circuit38, the base band circuit 300 need not supply the frequency set valuesof the IFVCO and TXVCO, and the determination of the usable bands of theIFVCO and TXVCO becomes possible in a shorter time.

And during the transmission, the control circuit 260 sends the frequencyset value for selecting the band of the TXVCO stored in the register 37to the arithmetic circuit 38, controls the arithmetic circuit 38 tocalculate the deviation of the KV value on the basis of the frequencyset value, determines the current correction value of the current sourceof the charge pump 237 a, which is necessary for correcting thedeviation, sends it to the charge pump 237 a, and corrects the current.Here, the current correction value sent to the charge pump 237 a in thiscase is the total of the correction value for correcting the deviationof the KV value due to the capacitance dispersions of the variablecapacitors Dv1, Dv2, and the correction value for correcting thedeviation of the KV value due to the capacitance dispersions of the bandswitching capacitors C11˜C42. Thus, the arithmetic circuit 38 maytotalize the correction values in real time; or it can totalize when thefrequency measurement value is obtained, and store the correction valueafter totalized in the register 37.

Next, the procedure of the frequency measurement of the IFVCO and TXVCOand the procedure of the correction of the KV value of the TXVCO in theradio communication system using the RF IC of this embodiment will beexplained with reference to FIG. 11. In FIG. 11, “Initial State”signifies the state after the power supply rises until thetransmission/reception becomes possible. “Dedicated Mode” signifies oneframe period of the state being possible of the transmission/receptionafter “Initial State”. Each frame is composed of eight slots. Thereception mode “Rx” is allocated to the first slot $0, and thetransmission mode “Tx” is allocated to the seventh slot $6. “Moni”allocated to the fifth slot $4 signifies the mode that measures thedistance from the base station while using a constant reception state asthe reference, and executes the measurement for determining the gain ofthe high gain amplifier during the reception. These modes are initiatedby the command that the base band circuit 300 supplies to the controlcircuit 260 of the RF IC 200. The command is configured with a code(called Word) having a predetermined bit length, and plural types ofcommand codes are prepared in advance.

The frequency measurement of the IFVCO and the frequency measurement ofthe TXVCO in the above embodiment are executed when a predeterminedcommand (Word 7) is inputted from the base band circuit 300 during theidle mode, such as the time slot “$7” in the “Initial State” being thetime slot for the GSM shown in FIG. 11, or the slot $0˜$3 and $7 in the“Dedicated Mode” in which both the transmission and the reception arenot executed. On the other hand, the selection of the band based on thefrequency measurement result and the correction of the KV value areexecuted at the start of the transmission mode “TX”. And In FIG. 11,“Rx” signifies the period while the reception burst appears after thereception mode command is transmitted.

After the power supply is on, when the base band circuit 300 suppliesthe RF IC 200 with the command named as “System Reset Word”, the radiocommunication system comes into the “Initial State”. Then, the controlcircuit 260 first starts the reference oscillator (VCXO) 264, andinitializes the circuits such as the registers inside the RF IC 200.Thereafter, as the command (Word 7) is inputted, the measurement of thecurrent Icp of the current source of the charge pump and the frequencymeasurement of the IFVCO and TXVCO are executed.

When the frequency measurement is competed, the base band circuit 300supplies the RF IC 200 with “Synthesizer Control Word” including thevalue (frequency information of usable channels) set to the counter 322.Then, the control circuit 260 selects the usable band of the RFVCO 250and sets the frequency value to the counter 322, on the basis of thefrequency information from the base band circuit 300 and the frequencymeasurement result stored in the register 37. And the control circuit260 puts the RFVCO 250 into the oscillation operation, and brings thePLL loop for reception into the lock-in state.

The “Synthesizer Control Word” includes a single control byte [TR] usedfor notifying the RF IC as to whether the next active slot is thetransmission slot or the reception slot. As the reception mode isselected, even if the “Synthesizer Control Word” is sent to the RF IC,the IFVCO and the IF synthesizer are not activated. The base bandcircuit 300 sends the command to designate the transmission mode, andthen it can send “Receive Word”; and when the RF IC receives the“Synthesizer Control Word”, the IFVCO and the IF synthesizer areactivated, and the oscillation frequency is locked in a correctfrequency to start the reception slot. And, when the “Receive Word” issent to the RF IC, the IFVCO and the IF synthesizer are automaticallydeactivated. On the other hand, as the transmission mode is demanded, tosend the “Synthesizer Control Word” to the RF IC will activate the IFVCOand the IF synthesizer. The PLL loop including the IFVCO is lockedbefore the transmission slot.

When the RF IC 200 receives “Receiver Control Word” to demand thereception operation from the base band circuit 300, the control circuit260 starts the offset cancel circuits inside the high gain amplifiers220A, 220B, to execute the input DC offset canceling of the amplifiers.After the DC offset canceling, the control circuit 260 enters thereception mode to put the reception system circuits into operation, andmakes the circuits amplify and demodulate the reception signal. And, thecontrol circuit 260 executes the switching control in answer to thereception signal for the GSM or the reception signal for the DCS/PCS.

When the reception mode is ended, the base band circuit 300 supplies theRF IC 200 with the command to demand the warm-up mode, “SynthesizerControl Word”, including the value (frequency information of usablechannels) being set to the counter 322 and the counter 323. Then, thecontrol circuit 260 enters the warm-up mode, selects the usable band ofthe RFVCO 250 and sets the frequency value to the counter 322 and thecounter 323, on the basis of the frequency information from the baseband circuit 300 and the frequency measurement result stored in theregister. And the control circuit 260 puts the RFVCO 250 and the IF VCO230 into the oscillation operation, and brings the RF PLL loop and theIF PLL loop into the lock-in state.

Thereafter, when the RF IC 200 receives “Transmitter Control Word” todemand the transmission operation from the base band circuit 300, thecontrol circuit 260 enters the transmission mode, selects the usableband of the TXVCO 240 on the basis of the frequency information from thearithmetic circuit 38 and the frequency measurement result stored in theregister 37, and operates the TXVCO 240 a or the TXVCO 240 b. Further,the control circuit 260 controls the conformable band determinationcircuit 39 to read out the current correction value corresponding to theselected band from the register 37 and send the current correction valueto the charge pump 237 a and correct the current Icp, and brings thetransmission PLL into the lock-in state to modulate and amplify thetransmission signal. And, the control circuit 260 turns on atransmission switch SW4, and executes the switching control of the TXVCOin answer to the transmission signal for the GSM or the reception signalfor the DCS/PCS. The use of either the TXVCO 240 a or the TXVCO 240 b isdetermined by a predetermined code contained in the command that thebase band circuit 300 supplies.

On the lower half of FIG. 11 is shown the detailed timing of thefrequency measurements of the IFVCO 230, TXVCO 240 a, and TXVCO 240 b,which are executed during the time slot “$7” of the “Initial State”. Asshown in FIG. 11, when the predetermined command “Word 7” is inputted,the measurement of the current Icp of the current source of the chargepump and the frequency measurement of the IFVCO are started. When thefrequency measurement of the IFVCO is ended, the frequency measurementof each band #0˜#14 of the TXVCO 240 a (GSM) and the frequencymeasurement of each band #0˜#14 of the TXVCO 240 b (DCS) aresequentially executed (period T2). Thereafter, in order to detect thedispersion of the KV value of the TXVCO, the frequency measurement(period T3) of band #15 of the TXVCO 240 a (GSM) at Vt=0 V and thefrequency measurement (period T4) of band #15 of the TXVCO 240 b (DCS)at Vt=2.8 V are sequentially executed.

And, the conformable band selection signals VB2′˜VB0′ determined on thebasis of these measurement values are supplied to the IFVCO at the startof the transmission mode; the conformable band selection signals VB3˜VB0and the current correction values of the charge pump determined on thebasis of the measurement values are supplied to the TXVCO at the startof the transmission mode. The frequency measurement value acquired bythe measurement for determining the conformable band being executedduring the period T2 at the control voltage Vt=1.0 V is also used as thefrequency measured for correcting the deviation of the KV value due tothe capacitance dispersions of the band switching capacitors C11˜C42.

Thus, the invention made by the inventors being described in detailbased on the embodiment, the invention is not limited to the abovedescriptions. In the above embodiment, for example, the frequencymeasurement of the TXVCO 240 a, TXVCO 240 b is carried out by using thecounter 322 and counter 323 prepared for the IFVCO 230, however themeasurement may be made by using the counter prepared for the RFVCO 250.And, instead of executing the frequency measurement of the TXVCO 240 a,TXVCO 240 b to all the bands, it may be arranged to execute themeasurement of only the odd bands or the even bands, and to acquire thefrequencies of unmeasured bands through calculating the averages of thefrequency measurement values of the bands before and behind. In theembodiment, the band number of the IFVCO 230 was 8, but it may be 4 or16 or the like.

The band number of the TXVCO 240 a, TXVCO 240 b is not limited to 16,and it may be 8 or 32 or the like.

Further, the above embodiment carries out the frequency measurement ofthe TXVCO 240 a, TXVCO 240 b by using the variable frequency divider 32prepared for the IFVCO 230, in view of restraining the circuit scalefrom increasing. However, it may be arranged to separately prepare avariable frequency divider (counter) for carrying out the frequencymeasurement of the TXVCO 240 a, TXVCO 240 b, and to carry out thefrequency measurement of the TXVCO 240 a, TXVCO 240 b in parallel to thefrequency measurement of the IFVCO 230, in response to one command. Toadd the above counter will slightly increase the circuit scale, but itwill finish the frequency measurement of plural VCOs in a shorter time.

Further in the description of the above embodiment, the count ‘1’ of thecorrection value for correcting the deviation of the KV value wascorrespondent to 1% of the current Icp of the current source of thecharge pump; however, the invention is not restricted to this. It doesnot matter at all whether the count ‘1’ of the correction valuecorresponds to 0.5% or 0.8% or the like of the current Icp of thecurrent source of the charge pump.

The above embodiment has been described mainly with the case in whichthe invention made by the inventors is applied to the RF IC used for theradio communication system of a mobile telephone capable of the EDGEsystem communication, which is the applicable field to background theinvention. However, the invention is not restricted to the RF IC of theEDGE system, but the invention can also be applied to a transmission VCOof the RF IC having the transmission system circuit that adopts theso-called polar loop system including the phase control loop and theamplitude control loop as the modulation method, in which the RF ICexecutes the modulation such as the 16 QAM or the 64 QAM for a radioLAN, requiring the phase modulation and the amplitude modulation.

1. A mobile communication device such as a mobile telephone, comprising:an antenna; an antenna switch which couples the antenna and switches atransmission signal and a reception signal; a communicationsemiconductor integrated circuit which couples the antenna switch andwhich demodulates the reception signal and modulates the transmissionsignal; a power amplifier which couples between the antenna switch andthe communication semiconductor integrated circuit device and whichamplifies a signal to be transmit from the communication semiconductorintegrated circuit device and outputs the transmission signal, whereinthe communication semiconductor integrated circuit device comprises: aphase control loop, including: a transmission oscillator which receivesa control voltage for determine a oscillation frequency of thetransmission oscillator and outputs a phase modulated signal, a phasecomparator which compares a phase of an output signal from thetransmission oscillator and a phase of a phase and amplitude modulatedsignal; and a charge pump which generates the control voltage accordingto an output of the phase comparator, and an amplitude control loop,including an amplitude comparator which compares an amplitude of thetransmission signal and an amplitude of a phase and amplitude modulatedsignal (TXIF) and which generates a voltage according to an amplitudedifference and outputs the voltage to the power amplifier as an outputlevel control signal, wherein a variation of the control voltage versusoscillation frequency characteristic of the transmission oscillator iscorrected by calibrating a current of the charge pump.
 2. A mobilecommunication device according to claim 1, wherein the transmissionoscillator is capable of an oscillation operation in a plural frequencyband.
 3. A mobile communication device according to claim 1, wherein thecurrent of the charge pump is calibrated in a manner that, when a valueof an oscillation frequency range divided by control voltage range ineach frequency band of the transmission oscillator is given by Kv, and acurrent value of the charge pump is given by Icp, a product value of Kvand Icp falls into a predetermined range.
 4. A mobile communicationdevice according to claim 3, the communication semiconductor integratedcircuit device further comprising: a frequency measurement circuit whichmeasures the oscillation frequency of the transmission oscillator; astorage circuit which stores frequency information measured in eachfrequency band of the transmission oscillator by the frequencymeasurement circuit; and an arithmetic circuit that calculates acorrection value of the current of the charge pump based on oscillationfrequency measurement values at two predetermined control voltages ofthe transmission oscillator.
 5. A mobile communication device accordingto claim 4, wherein each of the correction value of the current of thecharge pump correspond to the oscillation frequency measurement valuesare stored as table data in the storage means.
 6. A mobile communicationdevice according to claim 5, wherein the transmission oscillatorincludes a variable capacitors which are controlled by the voltagegenerated by the charge pump, plural set-capacitance capacitors, and aswitch circuit capable of connecting and disconnecting theset-capacitance capacitors, in which the switch circuit switches aconnection number of the set-capacitance capacitors to thereby vary theoscillation frequency band, and wherein the current correction values ofthe charge pump corresponding to the oscillation frequency bands eachare stored in the storage means.
 7. A mobile communication deviceaccording to claim 6, wherein the correction value includes a value tocorrect a variation of the Kv value due to a capacitance dispersion ofthe variable capacitor, and a value to correct a variation of the Kvvalue due to a capacitance dispersion of the set-capacitance capacitor.8. A mobile communication device according to claim 7, wherein thefrequency measurement of the transmission oscillator by the frequencymeasurement circuit is executed when the power supply is inputted andduring a period in which both a transmission operation and a receptionoperation are deactivated.
 9. A communication semiconductor integratedcircuit device according to claim 1, wherein the transmission oscillatorincludes a first oscillator that generates the transmission signal ofthe GSM system, and a second oscillator that generates the transmissionsignals of the DCS system and the PCS system.
 10. A mobile communicationdevice according to claim 9, the communication semiconductor integratedcircuit device further comprising: a third oscillator that generates asignal of an intermediate frequency, and a modulation circuit thatexecutes an orthogonal modulation to the signal of an intermediatefrequency generated by the third oscillator by means of I, Q signalsgenerated based on transmission data, wherein the phase modulated andamplitude modulated signal by the modulation circuit is supplied to thephase comparator and the amplitude comparator.
 11. A mobilecommunication device according to claim 10, wherein the frequencymeasurement circuit measures an oscillation frequency of thetransmission oscillator or the first oscillator and the secondoscillator, and an oscillation frequency of the third oscillator.